We discussed earlier that Pulse Amplitude Modulation (PAM) transmits information through amplitude scaling of the pulse according to the symbol value. To understand QAM, we have to differentiate between baseband and passband signals.
A baseband signal has a spectral magnitude that is nonzero only for frequencies around origin () and negligible elsewhere. An example spectral plot for a PAM waveform is shown below for 500 2-PAM symbols shaped by a Square-Root Raised Cosine pulse with excess bandwidth and samples/symbol . The PAM signal has its spectral contents around zero and hence it is a baseband signal.
For wireless transmission, this information must be transmitted through space as an electromagnetic wave. Also observe that once this spectrum is occupied, no other entity in the surrounding region can share the wireless channel for the purpose of communications. For these reasons (and a few others as well), a wireless system is allocated a certain bandwidth around a higher carrier frequency and the generated baseband signal is shifted to that specific portion of the spectrum. Such signals are called passband signals. That is why wireless signals in everyday use such FM radio, WiFi, cellular like 4G, 5G, and Bluetooth are all passband signals which execute their transmissions within their respective frequency bands sharing the same physical medium.
The easiest method to shift the spectrum to a designated carrier frequency is by multiplying, or mixing, it with a sinusoidal waveform. This is because time domain multiplication is frequency domain convolution between spectra of the sinusoid and the desired signal. Now in the article on DFT examples, we saw that the spectrum of a pure sinusoid at any frequency is composed of two impulses at that frequency (one positive and one negative and half amplitude). Also, convolution of a signal with a unit impulse results in the same signal shifted at the location of that impulse. Hence, a PAM waveform can be multiplied with a carrier sinusoid at frequency to “park” the actual spectrum in its allocated slot. This is drawn in Figure below.
After this spectral upconversion, both positive and negative portions of the baseband spectrum appear at and . The bandwidth — positive portion of the spectrum — hence becomes double. However, a relief comes from the fact that both and can be used to carry independent waveforms due to their orthogonality (having phases apart) to each other over a complete period, i.e.,
That is the birth of Quadrature Amplitude Modulation (QAM), in which two independent PAM waveforms are communicated through mixing one with a cosine and the other with a sine. Just like constellation, the term quadrature also comes from astronomy to describe position of two objects apart.
Next, we multiply the resulting complex signal with a complex sinusoid at carrier frequency and collect its real part, i.e., multiply arm with and arm with . The reason of including these two factors, and the negative sign, will be discussed later during QAM detection. Consequently, a general QAM waveform can be written as
where we have used the fundamental relation between continuous and discrete frequencies , and corresponds to the carrier frequency . Observe that is a real signal with no component. After digital to analog conversion (DAC), the continuous-time signal can be expressed as
In Eq (4), symbols determine the signal while control the signal. Such representation of the QAM waveform as a sum of amplitude scaled and pulse shaped sinusoids is known as the rectangular form. Using trigonometric identity , Eq (4) can also be written as
Called the polar form, here we have a single sinusoid whose amplitude and phase are determined by some combination of symbols and .
Some examples of QAM constellations are discussed below.
- For an even power of , square QAM is a constellation whose points are spaced on a grid in the form of a square. It is formed by a product of two -PAM constellations, one on axis and the other on axis. For example, a -QAM constellation is formed by two PAM constellations as drawn in Figure below, while some other square QAM constellations are also shown in the next Figure.
Notice how constellation points in higher-order QAM are closer to each other compared to lower-order QAM. A relatively lower noise power is then enough to cause a decision error by moving the received symbol over the decision boundary. This is the cost of increasing data rate by packing more bits in the same symbol. We will have more to say about it when we discuss Bit Error Rates (BER) for each constellation in another article.
For , the average symbol energy in square QAM is derived using Pythagoras theorem as
- A special case of -QAM is -PSK, which stands for Phase Shift Keying. As the name PSK implies, the amplitude remains constant in this configuration while the information is conveyed by different phases. Examples of -PSK, -PSK and -PSK are drawn in Figure below.
- -PSK constellation (also known as BPSK) looks similar to -PAM. However, the difference is that there is no carrier upconversion in PAM systems.
- -PSK constellation (also known as QPSK) is exactly the same as square -QAM.
- For , square QAM packs the constellation points more efficiently than and hence the modulation of choice in many wireless standards. Points come relatively closer in PSK and the closer the points, the larger the probability of receiving a symbol in error due to a jump across the decision boundary. On the other hand, QAM heavily depends on overall system linearity due to information conveyed in the signal amplitude. A constant envelope of the modulated signal is much more suitable for transmission over nonlinear channels where PSK is the preferred choice. In such situations, the performance of PSK is quite insensitive to nonlinear distortion and heavy filtering.
- Since any constellation that uses both amplitude and phase modulation fall under the general category of QAM, there can be many other rearrangements of QAM constellation points. However, we focus on square QAM and PSK in this text which are widely used in practice. Non-regular constellations can be formed which yield a small performance advantage, though it is usually not much significant. Moreover, the more points are in the constellation, the lesser is the difference.
To build a conceptual QAM modulator, we follow similar steps as in a PAM modulator. The block diagram for a -QAM modulator is drawn in Figure below.
- Every seconds, a new bit arrives at the input forming a serial bit stream.
- A serial-to-parallel (S/P) converter collects such bits every seconds that are used as an address to access two Look-Up Tables (LUT). One LUT stores symbol values while the other symbol values specified by the constellation.
- To produce a QAM waveform, the symbol sequences and are converted to discrete-time impulse trains in separate arms (one and the other ) through upsampling by , where is samples/symbol defined in as ratio of symbol time to sample time , or equivalently sample rate to symbol rate .
- As explained in sample rate conversion, upsampling inserts zeros between each symbol after which the interpolated intermediate samples can be raised from dead with the help of a pulse shaping filter in each arm that — in addition to shaping the spectrum — suppresses all the spectral replicas arising from upsampling except the primary one. These are the two and PAM waveforms forming the signal .
- Next, PAM waveform is upconverted by mixing with the carrier while the PAM waveform with , which are then summed to form the QAM signal . The carriers are generated through an oscillator at the Tx.
- This discrete-time signal is converted to a continuous-time signal by a DAC.
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The received signal is the same as the transmitted signal but with the addition of additive white Gaussian noise (AWGN) . The symbols are detected through the following steps illustrated in Figure below.
- Out of the infinite spectrum, the desired signal is selected with the help of a Bandpass Filter (BPF).
- Through an analog to digital converter (ADC), this signal is sampled at a rate of samples/s to produce a sequence of -spaced samples .
- Next, a complex signal is produced when by downconverted by mixing with two carriers, and which are generated through an oscillator at the Rx.
- The resulting complex waveform in and arms is processed through two matched filters thus generating and . As discussed earlier, the output of the matched filters are continuous correlations of the symbol-scaled pulse shape with an unscaled and time-reversed pulse shape.
- These and outputs are downsampled by at optimal sampling instants
to produce -spaced numbers and back from the signal.
- The minimum distance decision rule is employed in -plane to find the symbol estimates and to decide on the final constellation point.
Let us discuss the mathematical details of this process for a noiseless received signal as in Eq (4),
where is the carrier frequency and and are defined in Eq (1). After bandlimiting the incoming signal through a bandpass filter, it is sampled by the ADC operating at samples/second to produce
where the relation is used. To produce a complex baseband signal from the received signal, the samples of this waveform are input to a mixer which multiplies them with discrete-time quadrature sinusoids and yielding
Using the identities , and ,
Now we can observe why the two factors, a with both sinusoids and a negative sign with , were inserted.
- A at the modulator and later another in the detector result in a gain of , which cancels the halving of sinusoid amplitudes in above trigonometric multiplications.
- When a complex signal is multiplied with a complex sinusoid, a negative sign appears for phase addition of and in the section of cumulative waveform (see the multiplication rule for clarity). This section of the cumulative waveform is the real part of the complex product which is transmitted through the channel. Another negative sign with at the Rx delivers a positive at the output.
The matched filter output is written as
where comes into play from the definition of auto-correlation function. The double frequency terms in the above equation are filtered out by the matched filter , which also acts a lowpass filter due to its spectrum limitation in the range . To generate symbol decisions, -spaced samples of the matched filter output are required at . Downsampling the matched filter output generates
A square-root Nyquist pulse has an auto-correlation that satisfies no-ISI criterion,
In conclusion, the downsampled matched filter outputs map back to the Tx symbols in the absence of noise. If the world was simple, that would have been an end to it! But the world is complicated, and there are layers of issues that happen between the Tx information generation and Rx decision making. Anything we do after this will be to combat a subset of signal distortions and towards recovering the original information.
Observe that the system shown in Figure above is a multirate system. In the QAM detector, for example, the ADC and the matched filters operate at the sample rate . After the outputs of the matched filters are downsampled by , the symbol decisions are made at the symbol rate . Furthermore, there are some hidden assumptions in the QAM detector:
[Resampling] The ADC in general does not produce an integer number of samples per symbol, i.e., is not an integer. As we will see later, a resampling system is required in the Rx chain that changes the sample rate from the ADC rate to a rate that is an integer multiple of the symbol rate.
[Symbol Timing Synchronization] The peak samples at the end of symbol durations in both and arms are not known in advance at the Rx and in fact do not necessarily coincide with generated samples as well. This is because ADC just samples the incoming continuous waveform without any information about the symbol boundaries. This is a symbol timing synchronization problem which we will learn in Part II of the book.
[Carrier Frequency Synchronization] The carrier frequency of the oscillator at the Tx and that at the Rx are not exactly the same, equal to . Instead, if the oscillator frequency at the Tx is denoted as , then at the Rx, we have , where is the difference between the two and can be either positive or negative. Any movement by the Tx, the Rx or within the environment between them also causes a shift in frequency (known as Doppler shift) that needs to be compensated. We will discuss carrier frequency synchronization in Part II of the book.
[Carrier Phase Synchronization] Furthermore, the carrier phase at the Rx oscillators is not known beforehand and needs to be estimated which will be explained in Part II of the book.
QAM Eye Diagram and Scatter Plot
As discussed above, a QAM signal consists of two PAM signals riding on orthogonal carriers. At baseband, these two PAM signals appear as and components of a complex signal at the Tx and Rx. Therefore, there are two eye diagrams for a QAM modulated signal, one for and the other for and both of them are exactly the same as PAM.
The case of scatter plot is a little different. After downsampling the matched filter output to symbol rate, the samples thus obtained are mapped back to the constellation, previously illustrated in QAM detector block diagram and now drawn in Figure below for -QAM modulation. This cloud of samples around the constellation points is now -dimensional (for PAM, there was no component) and can also be understood as a plot of versus for each mapped value.
Looking at the scatter plot, one can readily deduce a lot of features for the particular transmission system. At this stage, however, it is enough to observe that the diameter of these clouds is a rough measure of the noise power corrupting the signal. For the ideal case of no noise, this diameter is zero and all the optimally timed samples coincide with their respective constellation points.